Commutation cells are commonly used in electronic systems that require conversion of a voltage source, including both DC-DC and DC-AC converters. These commutation cells are based on the use of power electronic switches, for example metal-oxide-semiconductor field-effect transistors (MOSFET). FIG. 1 is a schematic representation of a MOSFET. The shown MOSFET 10 has a Drain, a Gate and a Source and is controlled by signals 12 applied by a gate driver 14 connected to the Gate via a gate resistor RG. As is well-known, the MOSFET 10 includes parasitic (or stray) capacitances such as CGD between the Drain and the Gate, CGS between the Gate and the Source, and CDS between the Drain and the Source. A sum of the capacitances CGD, CGS and CDS is often referred to as an output capacitance COSS of the MOSFET 10. A parasitic diode Dp is present between the Drain and the Source. While current may flow in the MOSFET 10 from the Drain to the Source, current may flow in the diode Dp in the reverse direction. A connection of the MOSFET 10 to a circuit creates parasitic inductances LD at its drain and LS at its source. The presence of parasitic components of the MOSFET 10 greatly impacts its behavior upon switching. The MOSFET 10 is well known to those of ordinary skill in the art and is not described further herein.
FIG. 2 is a circuit diagram of a conventional MOSFET-based DC-DC converter. In the circuit 20, power from a source 22 having a voltage Vin is converted to another DC voltage applied to a load (shown as LOAD on FIG. 2). The LOAD may consist of a purely resistive element or may also include capacitive and/or inductive components. A capacitor Cout placed in parallel with the LOAD and an inductor Lout placed in series with the LOAD form a filter that averages a voltage across an opposite diode Do, which is described hereinbelow, so that a voltage on the LOAD remains fairly constant.
The circuit 20 includes a commutation cell and a control component (described in details hereinbelow) and controls the voltage applied to the LOAD by the source 22. The commutation cell of the circuit 20 comprises a main switch Qm, which may for example consist of a MOSFET or like power electronic switch, controlled by a gate driver (not shown but shown on FIG. 1), and the opposite diode Do. The commutation cell further includes an equivalent source capacitor (not shown) in parallel with the source 22 and an equivalent current source (not specifically shown) for a current Iout that flows in the LOAD.
When the main switch Qm is open, the current Iout flows from the LOAD through the opposite diode Do and returns to the LOAD, as reflected by the arrows 202 and 204. When the main switch Qm is closed, the current Iout flows through the main switch Qm and the source 22 and returns to the LOAD, as reflected by the arrows 206, 208 and again 204. When the main switch Qm is closed, a voltage between its drain and source is zero (or substantially zero) and the entire voltage Vin, for example 450 Volts DC, is applied across the opposite diode Do. It is however intended to use the circuit of FIG. 2 to apply a controlled DC voltage to the LOAD, this controlled DC voltage being lower than the voltage Vin of the source 22. To this end, the commutation cell is switched on and off at a rapid rate, a duty cycle of the commutation cell controlling an effective voltage applied to the LOAD. The voltage applied on the LOAD is equal to the duty cycle multiplied by the voltage Vin of the source 22.
The auxiliary components of the circuit 20 comprise an auxiliary capacitor Caux, an auxiliary inductor Laux, diodes D1, D2 and D3, as well as an auxiliary switch Qa. Initially, when the main switch Qm is conducting the entire current Iout (see arrow 206) the voltage between its drain and source is zero. At that time, a voltage on the auxiliary capacitor Caux is substantially equal to Vin. Opening the main switch Qm, the current Iout is gradually deviated in the auxiliary capacitor Caux, arrow 210. Accordingly, the voltage slope is limited and therefore allows Qm to turn-off at almost zero voltage, thereby reducing switching losses in the main switch Qm.
Then, while the main switch Qm is off, the auxiliary switch Qa, for example another MOSFET, is closed in preparation for closing the main switch Qm again. The voltage that was applied across the auxiliary switch Qa is gradually deviated across the auxiliary inductor Laux. Accordingly, the current slope is limited and therefore allows Qa to turn-on at almost zero current, thereby reducing switching losses in the auxiliary switch Qa.
A portion of the current Iout is now directed via the auxiliary inductor Laux to this auxiliary switch Qa, see arrows 212 and 214. The auxiliary inductor Laux limits the voltage across the auxiliary switch Qa in order to minimize switching losses. As a voltage across the auxiliary switch Qa decreases, while it becomes closed, its current variation di/dt increases, causing a voltage across the auxiliary inductor Laux to also increase. At the time when the auxiliary switch Qa becomes fully conductive (i.e. when it is fully closed), the voltage Vin is substantially present on the auxiliary inductor Laux and the current variation di/dt becomes equal to Vin divided by the value of the auxiliary inductor Laux. Until this moment, at least a part of the current Iout was flowing through the opposite diode Do, along arrow 202, on which a voltage is initially at or near zero. When the entire current Iout flows through the auxiliary switch Qa, arrow 214, a recovery current starts flowing in reverse direction in the opposite diode Do, in a direction opposite to that of arrow 202. This recovery current in the opposite diode Do has a rate of Vin divided by Laux. Rapidly, once charges accumulated on the PN junction of the opposite diode Do are withdrawn, the opposite diode Do becomes blocked and a resonance is initiated between the auxiliary inductor Laux and parasitic capacitances (not shown) of the opposite diode Do, of the main switch Qm and of another diode D2 through the auxiliary capacitor Caux. It is observed that the auxiliary capacitor Caux is much larger than the parasitic capacitance of the diode D2. Energy accumulated in these capacitors is transferred to the auxiliary inductor Laux after a quarter of a resonant cycle.
Because the opposite diode Do is now blocked and because a resonance with the above mentioned capacitors has taken place for quarter of a cycle, a voltage thereacross is equal to Vin. A voltage on the main switch Qm is therefore zero, allowing closing of this main switch Qm at zero voltage. The auxiliary switch Qa is then opened in order to limit losses in the auxiliary inductor Laux. The current that was flowing in the auxiliary switch Qa is gradually deviated to the auxiliary capacitor Caux. Accordingly, the voltage slope is limited and therefore allows Qa to turn-off at almost zero voltage, thereby reducing switching losses in the auxiliary switch Qa, thereby reducing switching losses in the auxiliary switch Qa. As expressed hereinabove, charges accumulated on the auxiliary capacitor Caux will be transferred to the main switch Qm when this latter switch opens, in another cycle. No significant energy is spent in the auxiliary capacitor Caux. This current in the auxiliary inductance reaches zero after a time dictated by the current that flows therein upon opening of switch Qa and a current variation in the auxiliary inductance di/dt that defined as the source voltage Vin divided by the auxiliary inductor Laux.
Upon opening of the auxiliary switch Qa, due to the current variation di/dt in the diode D1 at a rate of Vin divided by Laux, there will be a recovery current in the diode D1, which is in series with the auxiliary inductor Laux. Another recovery current will also be present in the diode D3, but it will be very small because the auxiliary capacitor Caux is large. The voltage across the diode D3 remains near zero. Yet another recovery current in the diode D2 will be very small because an output capacitance COSS of the auxiliary switch Qa is large compared to the parasitic capacitance of the diode D1 and further because the diode D2 is smaller than the diode D1. These recovery currents are however of secondary importance because the diodes D2 and D3 are much smaller than the opposite diode Do. The diode D1 provides a soft recovery because it is larger than the diodes D2 and D3 and because, as mentioned hereinabove it turns off with the current variation di/dt at a rate of Vin divided by Laux.
Those familiar with the circuit 20 will appreciate that it can be operated at a duty cycle in a range from zero to 100 percent, wherein the duty cycle is defined as a ratio of the closing time of the switches over a complete cycle time of the commutation cell. It is however required to allow full closing of the switches when the duty cycle is greater than zero. It is also required to prevent opening of the switches if the duty cycle is so great that the auxiliary capacitor Caux is prevented from fully discharging upon opening of the main switch Qm.
The circuit 20 of FIG. 2 suffers from the following drawbacks.
Firstly, the recovery current in the opposite diode Do, which is a large diode and, in case of a full leg, the very slow parasitic diode of a MOSFET, generates significant losses occurring in the circuit 20, primarily in the diode itself and in the auxiliary inductor Laux as well as in the auxiliary switch Qa due to the recovery current before added to the load current. These losses impact the efficiency of the circuit 20 and limit the switching frequency. The losses generate heat that must in most cases be dissipated, cumulated with the limited switching frequency, which in turn impact the physical size of a converter built on the basis of the circuit 20.
Secondly, the resonance between various elements of the circuit 20, in particular at the time of blocking the opposite diode Do, when the recovery current causes a very high dV/dt across the parasitic capacitance of the opposite diode Do, causes significant noise in terms of electromagnetic interference (EMI). This EMI may be detrimental to many applications and may require complex filtering, which in turn may involve an increase of the size and/or cost of the physical implementation of the circuit 20, or a decrease of its performance, or both.
Finally, the energy flow may only circulates from the high voltage to the low voltage (unidirectional from input to output) so it can only be used has a DC/DC buck converter.
Therefore, there is a need for improvements to converter circuits that compensate for problems related to poor efficiency and electromagnetic noise generation in power electronics circuits and to improve the flexibility to allows the operation as DC/DC boost converter, as DC/AC converter or as AC/DC converter.